Electronic circuit for tripling frequency

ABSTRACT

In an embodiment, a circuit for tripling frequency is configured to receive an input voltage (V in ) having a sinusoidal shape and a base frequency. The circuit has a first and a second transistor pair that are cross-coupled, and a trans-characteristics f(V in ) approximating a polynomial nominal trans-characteristic given by 
     
       
         
           
             
               f 
                
               
                 ( 
                 
                   V 
                   in 
                 
                 ) 
               
             
             = 
             
               
                 ( 
                 
                   
                     
                       3 
                       A 
                     
                      
                     
                       V 
                       in 
                     
                   
                   - 
                   
                     
                       4 
                       
                         A 
                         3 
                       
                     
                      
                     
                       V 
                       in 
                       3 
                     
                   
                 
                 ) 
               
                
               
                 g 
                 m 
               
             
           
         
       
     
     where A represents an amplitude of the input voltage and g m  is a transconductance of transistors of the first and second transistor pairs.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of Italian Patent Application No.102019000016871, filed on Sep. 20, 2019, which application is herebyincorporated herein by reference.

TECHNICAL FIELD

The present invention relates to an electronic circuit, and inparticular embodiment, to an electronic circuit for tripling frequency.

BACKGROUND

As is known, communications at millimeter-wave (mm-wave) range havedrawn a lot of attention in recent years due to the wide availablebandwidth yielding higher data transmission capacity. Thus, currentsystems use transceivers that convert the exchanged signals from thebase frequency to the selected communication frequency and vice versa.To this end, the transceivers use circuitry to generate a localoscillation (LO). Design of the local oscillation transceivers iscritical because many conflicting parameters, i.e., tuning range, phasenoise, output power and level of spurious tones, affect theperformances. Differently from what is commonly pursued at other radiofrequencies, local oscillation generation with a PLL (Phase Locked Loop)circuit comprising a VCO (Voltage Controlled Oscillator) at the desiredoutput frequency is not viable at mm-wave range. In fact, the severeimpact of parasitic structures and effects in silicon technology and thelow quality factor of passive components (mostly, variable capacitors)impair the achievable tuning range and phase noise. Moreover,traditional frequency dividers in the PLL cause excessive powerconsumption.

A more promising approach consists in providing a PLL in a lower range(e.g., in the 10-20 GHz range), where the silicon VCO features the bestfigure of merit, followed by a frequency multiplier chain.

For example, FIG. 1 shows a typical frequency multiplier system 1 forobtaining a multiplication by 6, for example, for obtaining a 60 GHzvoltage from a 10 GHz source. Here, a low-frequency voltage generator 2supplies an input voltage Vin at a base frequency f_(o), e.g., at 10GHz, to a frequency tripler 3, which generates an intermediate voltageV1 at a triple frequency (3f_(o)). The intermediate voltage V1 isavailable at a first output O1 through a first buffer 4 and fed to afrequency doubler 5 that generates an output voltage Vo at outputfrequency 6f_(o). The output voltage Vo is made available at a secondoutput O2 through a second buffer 6.

The frequency multiplier system 1 has to provide a good suppression ofthe driving signal and of undesired harmonics in order not to impair thetransceiver performance. In particular, in the frequency multipliersystem 1, it is desired that the first stage (frequency tripler 3)features the highest suppression, because its spurious tones are shiftedclose to the final LO frequencies by the intermodulation of the cascadedstages. Moreover, this issue is more critical for odd-order multipliers,because even-order multipliers (here, the frequency doubler 5) mayexploit push-push transistors for suppression of signal components atundesired frequencies.

Odd-order multipliers, such as the frequency tripler 3 of the frequencymultiplier system 1, typically comprise a transistor with low conductionangle (e.g., class-C biased transistor) that generates a harmonic-richcurrent and the desired component is selected with a band-pass filter oran injection-locked oscillator.

For example, FIG. 2 shows the basic structure of a class-C triplercircuit, indicated by 10. The class-C tripler circuit 10 comprises atransistor 11, here of bipolar type, fed at a base terminal B by theinput voltage

Vin=A sin(2πf _(o) t)

through an input capacitor 12 and coupled to a bias voltage Vb through aresistor 13. The transistor 11 has an emitter terminal E grounded, and acollector terminal C coupled to an output terminal 14 and to a supplyvoltage VCC through an LC resonant circuit 15 tuned at 3f_(o).

In a per se known manner, the class-C tripler circuit 10 conducts acurrent Io whose harmonic content is set by the conduction angle θdetermined by the bias voltage Vb and shown in the simulations of FIGS.3A and 3B.

In detail, FIG. 3A shows the plot of the output current density Jo(output current Io normalized to the area A of the transistor 11) forthe fundamental (If_(o)), the third harmonic (I3f_(o)) and the fifthharmonic (I5f_(o)). FIG. 3B shows the plot of the ratio If_(o)/I3f_(o)(=Jf_(o)/J3f_(o)) of the amplitudes of the normalized fundamental Jf_(o)and the normalized third harmonic (J3f_(o)) as a function of theconduction angle θ.

As may be seen in FIG. 3A, the total harmonic rejection ratio HRR isdominated by the fundamental Jf_(o), i.e., the leakage of the drivingsignal, which is always larger than the normalized third harmonicJ3f_(o). At conduction angle θ≈150°, the amplitude of normalized thirdharmonic J3f_(o) is maximized (point M in FIG. 3A), but the fundamentalJf_(o) is still 9 dB larger than the third harmonic J3f_(o) (as alsovisible in FIG. 3B). Even dimensioning the LC resonant circuit 15 tomaximize suppression while compromising bandwidth, the class-C triplercircuit 10 has a very poor suppression, that cannot be increased over 20dB.

Class-C tripler circuits may be improved, in principle, by using a morecomplex filter topology or by cascading multiple filtering stages, butat the cost of a high design complexity, big area, bandwidth limitationand higher consumption.

Also the injection-locked oscillator solution (see, e.g., N. Mazor etal., “A high suppression frequency tripler for 60-GHz transceivers,” in2015 IEEE MTT-S International Microwave Symposium, 2015, pp. 1-4),although providing a better suppression (up to about 30 dB), does notsatisfactorily solve the problem.

SUMMARY

Some embodiments provide a frequency tripler that improves thesuppression of the driving signal frequency at the output.

Some embodiments relate to tripling frequency, in particular forradiofrequency applications in the millimeter-wave range.

Some embodiments relate to an electronic circuit for tripling frequency.Some embodiments related to a corresponding method.

BRIEF DESCRIPTION OF THE DRAWINGS

For the understanding of the present invention, embodiments thereof arenow described, purely as a non-limitative example, with reference to thedrawings, wherein:

FIG. 1 shows a block diagram of the general structure of a frequencymultiplier chain;

FIG. 2 is a circuit diagram of a known frequency multiplier;

FIGS. 3A and 3B are plots of quantities related to the circuit of FIG.2;

FIG. 4 is a circuit diagram of a frequency tripler, according to anembodiment of the present invention;

FIG. 5 shows the plot of a desired trans-characteristic and the actualcharacteristic of the circuit of FIG. 4, according to an embodiment ofthe present invention;

FIG. 6 is a circuit diagram of a tripler device including the circuit ofFIG. 4, according to an embodiment of the present invention;

FIG. 7 is a circuit diagram of an embodiment of a block of the triplerdevice of FIG. 6;

FIG. 8 shows plots of quantities related to the circuit of FIG. 4,according to an embodiment of the present invention; and

FIG. 9 shows comparative plots of the power spectrum obtained with thecircuit of FIG. 4 and the circuit of FIG. 1.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

FIG. 4 shows a tripler circuit 20 that allows obtaining a highsuppression of undesired harmonics, according to an embodiment of thepresent invention.

The tripler circuit 20 represents an implementation of an idealtransistor-based tripler circuit having a polynomialtrans-characteristic f(V_(in)) (later on also called ideal polynomialtrans-characteristic) according to the following equation (1):

$\begin{matrix}{{f\left( V_{in} \right)} = {\left( {{\frac{3}{A}V_{in}} - {\frac{4}{A^{3}}V_{in}^{3}}} \right)g_{m}}} & (1)\end{matrix}$

where g_(m) is the transconductance of the transistor in the triplercircuit (at a specific DC biasing condition).

In particular, as may be demonstrated with some calculation, the aboveideal trans-characteristics allows a tripler circuit, receiving at itsinput a sinusoidal driving voltage:

V _(in) =A sin(2πf ₀ t)=A sin(ω₀ t)

having amplitude A and base frequency f_(o) is able to generate anoutput current Io:

I _(o) =f(V _(in))=g _(m) sin(32πf ₀ t)=g _(m) sin(3ω₀ t)

having only the third harmonic (3f_(o)).

FIG. 4 shows the structure of the tripler circuit 20 that has atrans-characteristics approximating the above polynomialtrans-characteristics f(V_(in)), as discussed later on.

In detail, with reference to FIG. 4, the tripler circuit 20 comprises afirst and a second pair of transistors, cross-coupled to each other. Inparticular, the first pair of transistors comprises a first and a secondtransistor Q1, Q2, here of the bipolar NPN-type, and the second pair oftransistors comprises a third and a fourth transistor Q3, Q4, here alsoof the bipolar NPN-type. Transistors Q1-Q4 have same parameters, inparticular same emitter area.

In detail, the first and second transistors Q1, Q2 have emitterterminals coupled to each other and to a common node 21, base terminalscoupled to a first and, respectively, a second input node 22, 23 andcollector terminals coupled to a first and, respectively, a secondoutput node 24, 25 supplying a first and, respectively, a secondsingle-ended current Io− and Io+.

The third and fourth transistor Q3, Q4 have emitter terminals coupled toeach other and to the common node 21, base terminals coupled to a thirdand, respectively, a fourth input node 27, 28, and collector terminalscoupled to the second and, respectively, the first output node 25, 24.

A biasing current source 26, configured to generate bias current I_(b),is coupled between the common node 21 and ground.

The first and the second input nodes 22, 23 receive each a fractionequal to ½ of the input voltage V_(in), in counter-phase, both reducedby a DC voltage (offset voltage V_(os)). The third and the fourth inputnodes 27, 28 receive each an attenuation α/2 of the input voltage, incounter-phase, the attenuation α being selected so that, duringoperation, at low values of the input voltage V_(in) and consideringalso the offset voltage V_(os), the first pair of transistors Q1, Q2 isstill off, while the second pair of transistors Q3, Q4 are on, asdiscussed in detail below.

Specifically, the first input node 22 receives first voltage V1:

V1=V _(in)/2−V _(os);

the second input node 23 receives voltage V2:

V2=−V _(in)/2−V _(os);

the third input node 27 receives voltage V3:

V3=αV _(in)/2; and

the fourth input node 28 receives voltage V4:

V4=−αV _(in)/2

where V_(os) is the DC offset voltage and α is the attenuation, asindicated above.

The tripler circuit 20 of FIG. 4 operates as follows. As indicatedabove, at small values of the input voltage V_(in), the first and secondtransistors Q1, Q2 have low base-to-emitter biasing voltages and areoff; thus the output currents I_(o+) and I_(o−) are governed only by thethird and fourth transistors Q3, Q4, approximating equation (i). Whenthe value of the input voltage V_(in) increases, the first and secondtransistors Q1, Q2 turn on, subtracting current from the output nodes24, 25. In particular, after reaching their maximum amplitudes, theoutput differential current I_(o)=I_(o+)−I_(o+) reduces, reversing theslope of the trans-characteristic.

The trans-characteristic of the tripler circuit 20 of FIG. 4 is shown bycurve A plotted in FIG. 5 with dotted line as normalized output currentI_(on) versus normalized input voltage V_(in)/A. For reference, FIG. 5shows also the ideal trans-characteristic (i) with continuous line B.

In particular, the normalized output current I_(on) is the differentialcurrent I_(o+)−I_(o+), normalized with respect to its maximum amplitude(equal to I_(b)).

The values of attenuation α and offset voltage V_(os) are selected sothat the trans-characteristic of the tripler circuit 20 tracks the idealpolynomial trans-characteristic of equation (1), that is so that thetrans-characteristic of the tripler circuit 20 is null at V_(in)=0, thenincreases with a similar slope as the ideal polynomialtrans-characteristic, then decreases again to zero and to negativevalues, following the ideal trans-characteristic. The opposite happensfor negative values of V_(in).

In particular, the zero-crossings (besides of that at V_(in)=0) of thetrans-characteristic of the tripler circuit 20 occur when the voltage atthe base terminal of the third transistor Q3 (at the third input node27) equals the voltage at the base terminal of the first transistor Q1(at the first input node 22) as well as when the voltage at the baseterminal of the fourth transistor Q4 (at the fourth input node 28)equals the voltage at the base terminal of the second transistor Q2 (atthe second input node 23), that is when condition (2) is satisfied:

$\begin{matrix}{{{{\pm \frac{V_{in}}{2}} - V_{os}} = {{\pm \alpha}\frac{V_{in}}{2}}}.} & (2)\end{matrix}$

The zero-crossings occur thus at the following values of the inputvoltage V_(in):

$\begin{matrix}{{V_{in} = {\pm \frac{2V_{os}}{\left( {1 - \alpha} \right)}}}.} & (3)\end{matrix}$

On the other hand, the zero-crossings of the trans-characteristics (1)(besides of that at V_(in)=0) occur at the following values of the inputvoltage V_(in):

$\begin{matrix}{{V_{in} = {{\pm \frac{\sqrt{3}}{2}}A}}.} & (4)\end{matrix}$

It follows that the trans-characteristic of the tripler circuit 20 andthe ideal trans-characteristic have same zero-crossings when attenuationα and offset voltage V_(os) satisfy the following condition:

$\begin{matrix}{{\frac{2V_{os}}{\left( {1 - \alpha} \right)} = {\frac{\sqrt{3}}{2}A}}.} & (5)\end{matrix}$

Analysis of the derivatives of the ideal polynomial trans-characteristic(i) shows that its slope at the zero crossings at V_(in)=±√{square rootover (3)} A/2 is ±2 times that of the origin. Further circuit analysisproves that it is enough to design attenuation α=0.2 to have the slopesof the two trans-characteristics identical at zero crossings, such thatthe shape of the actual trans-characteristic of the tripler circuit 20keeps as close as possible to the ideal one (see FIG. 5). However,circuit analysis proves that a value of attenuation α comprised in therange 0.1 to 0.35 allows the trans-characteristic of the tripler circuit20 to suitably fit the plot of the ideal polynomial trans-characteristic(1). In fact, the spread of the value of attenuation α may becompensated through the offset voltage V_(os), as discussed below.

By fixing the value of attenuation α, the value of the offset voltageV_(os) is obtained as a linear function of the amplitude A of the inputvoltage V_(in), based on condition (5).

In this case, also a non-optimal value of the attenuation α may be set,and the envelope detector operates as an open loop able to compensateand maintain the linear desired relationship of condition (5).

For example, FIG. 6 shows a tripler circuit, indicated by 30, comprisingthe tripler 20 of FIG. 5 and an envelope detector, according to anembodiment of the present invention.

In FIG. 6, an input transformer T1 has a primary winding 31 coupledbetween a first and a second circuit input 32, 33 and a secondarywinding 35 coupled between the input nodes 22, 23 of the tripler circuit20. The secondary winding 35 has a central tap 36 connected to a firstoutput 37 of an envelope detector 38 and set at a first biasing voltageVb1.

A voltage divider 40, of capacitive type, is coupled between the inputnodes 22, 23 of the tripler circuit 20 and comprises a first branch 41and a second branch 42.

The first branch 41 of the voltage divider 40 comprises a firstcapacitor 45, a first resistor 46, a second resistor 47 and a secondcapacitor 48 connected in series. The first and second capacitors 45, 48have same capacitance C1; the first and second resistors 46, 47 havesame resistance R.

The first branch 41 has a central tap between the first and secondresistors 46, 47 coupled to a second output 50 of the envelope detector38, which generates a second biasing voltage Vb2. The first branch 41also has a first intermediate node 51 between the first capacitor 45 andthe first resistor 46 and a second intermediate node 52 between thesecond resistor 47 and the second capacitor 48. The voltage differenceVb2-Vb1 forms the offset voltage V_(os) of the tripler 20 of FIG. 5.

The second branch 42 of the voltage divider 40 comprises a thirdcapacitor 54 coupled between the first and second intermediate nodes 51,52. The third capacitor 54 has a capacitance C2. First and secondintermediate nodes 51, 52 are also coupled to the third and,respectively, the fourth input node 27, 28 of the tripler circuit 20.

The first and second input nodes 22, 23 of the tripler circuit 20 arealso coupled to a first, respectively a second input 55, 56 of theenvelope detector 38 through respective capacitors 57, 58.

The tripler circuit 30 also comprises an output transformer T2 having aprimary winding 61 coupled between the first and second output nods 24,25 of the tripler 20 and a second winding 62 coupled between a first andsecond circuit outputs 64, 65; and an LC network 66 formed by a shuntcapacitor 67 and a tail inductor 68 is coupled between the common node21 of the tripler 20 and ground.

The first and second circuit outputs 64, 65 may be connected to anoutput buffer similar to the first buffer 4 of frequency multipliersystem 1 of FIG. 1 and/or to a frequency multiplier such as thefrequency doubler 5 of the frequency multiplier system 1 of FIG. 1.

In the tripler circuit 30 of FIG. 6, the first transformer T1 operatesfor line adaptation (as a balun transformer) and generates adifferential signal (corresponding to input voltage V_(in) of FIG. 4 andthus identified with the same reference) directly applied on the firstand second input nodes 22, 23 of the tripler circuit 20 (and thus on thebase terminals of the first and second transistors Q1, Q2). Thedifferential signal V_(in) is reduced by the attenuation α by thecapacitive divider 40 and applied to the third and fourth input nodes27, 28 of the tripler circuit 20 (and thus on the base terminals of thethird and fourth transistors Q3, Q4).

The tail inductor 68 resonates with shunt equivalent capacitanceexisting at the common node 21 and the shunt capacitance 67 is sizedsufficiently large to act as an AC-short at the operating frequency ofcommon node 21, which is 2f_(o). In fact, LC network 66 allows the shuntparasitic capacitances at common node 21 to charge-discharge at highfrequency using the current being exchanged with the tail inductor 68,hence not to lag behind the base voltages of the input transistors Q1-Q4when operating at high input frequency.

FIG. 7 shows an exemplary implementation of envelope detector 38generating the offset voltage V_(os) that satisfies condition (5),according to an embodiment of the present invention.

In the specific implementation shown in FIG. 6, envelope detector 38comprises an input differential pair 70 formed by a fifth and a sixthtransistor Q5 and Q6 and driven by the input voltage V_(in). A currentsource 71, generating a second reference current 2I_(REF), twice thereference current I_(REF), is coupled to the emitter terminals of thefifth and sixth transistors Q5 and Q6 and to an averaging RC filter 72.The averaging filter RC 72 includes an averaging resistor 73 havingresistance R_(E) and is coupled to a first voltage generating network75. First voltage generating network 75 comprises a first currentgenerating branch 76 and a first current mirroring branch 77. Firstcurrent generating branch 76 includes a first current source 80generating reference current I_(REF) and a transistor Q7; first currentmirroring branch 77 includes a transistor Q8 that is base-coupled totransistor Q7 of the first current generating branch 76, has a collectorterminal forming first output 37 of the envelope detector 38 andgenerates the first biasing voltage Vb1. A second voltage generatingnetwork 81, having the same basic structure of the first voltagegenerating network 75, has coupled transistors Q9 and Q10 and generatesthe second biasing voltage Vb2 at a collector of transistor Q9 coupledto the second output 50 of the envelope detector 38. The second voltagegenerating network 81 also includes a second current source 82generating the reference current I_(REF).

A supply voltage V_(CC) is applied to the collector terminals of thefifth and sixth transistors Q5 and Q6 and to supply nodes of the firstand second voltage generating networks 75, 81. Supply voltage V_(CC) isalso applied to a central tap of the second transformer T2.

All transistors Q5-Q10 in the envelope detector 38 share a same biasvoltage V_(CM). In this way, transistors Q5-Q6 (driven by |V_(in)(t)|)and Q7, cause the voltage V_(RE) on averaging resistor 73 to be equal tothe average value of |V_(in)(t)|. Since V_(in)(t)=A sin(2πf_(o)t),voltage on the averaging resistor 73 is V_(RE)=(4/π)A and the currentthrough it is I_(RE)=(4/π)A/R_(E). MOSFET transistors M1, M2 in thefirst voltage generating network 75 mirror current I_(REF)+I_(RE) into afirst output resistor 85 (coupled to the first output 37 of the envelopedetector 38) with resistance while MOSFET transistors M3, M4 in thesecond voltage generating network 81 mirror current I_(REF) into asecond output resistor 86 with resistance R2 (coupled to the secondoutput 50 of the envelope detector 38). Thus:

Vb1=V _(CC)−(I _(REF) +I _(RE))·R1,

Vb2=V _(CC) −I _(REF) ·R2.

Assuming R1=R2,

V _(os) =Vb2−Vb1=R1·I _(RE)=(4/π)(R1/R _(E))A.

The ratio R1/R_(E) is designed such that V_(os) satisfies condition (5),thus allowing to maintain good suppression of the fundamental frequencycomponent independently from the amplitude of the input signal.

Measurements made by the Applicant confirm that the tripler circuit 20suppresses almost completely the component at fundamental frequencyf_(o) in the output current I_(o). For example, FIG. 8 shows the plot ofthe output power P_(3fo) measured at 3f_(o) when the input power P_(in)is swept at 1.2.5 GHz and the plot of the sum P_(sum) of the outputpowers P_(fo) (measured at f_(o)) and P_(5fo) (measured at 5f_(o))obtained with the frequency tripler 20. As may be seen, in the range −5dBm to 10 dBm, the output power P_(3fo) is higher than the sum powerP_(sum) of about 40 dBm in almost the entire input power range and inany case never lower than 36 dBm.

The improvement of the tripler circuit 20 with respect to a conventionaltripler using class-C operating transistors is also visible from FIG. 9,showing the output spectrum obtainable with the present frequencytripler 20 (curve Tr1_3f0), the undesired output spectrum at f_(o)obtainable with a conventional tripler using class-C operatingtransistors (curve Tr2_f0) and the undesired output spectrum at f_(o)obtainable with the present frequency tripler 20 (curve Tr1_f0). Asvisible, curve Tr1_f0 is 20 dB lower than curve Tr2_f0 and 40 dB lowerthan curve Tr1_3f0.

Advantages of embodiments of the present invention are clear from theabove. For example, it is underlined that, in some embodiments, thetripler circuit is advantageously able to suppress undesired fundamentaland harmonics in a much better way than with conventional circuits.

In some embodiments, the tripler circuit advantageously operates at lowpower compared with conventional designs exploiting class-C transistorsand filters.

Finally, it is clear that numerous variations and modifications may bemade to the frequency tripling electronic circuit described andillustrated herein, all falling within the scope of the invention asdefined in the attached claims.

For example, the bipolar transistors Q1-Q10 could be replaced by MOSFETtransistors; the transistors may be made in any technology, such assilicon, gallium arsenide (GaAs), indium phosphide (InP), etc.; thestructure of the envelope detector may be any other, provided itperforms the functionalities depicted above, specifically it getsVb2−Vb1=Vos satisfying relation (5).

What is claimed is:
 1. A circuit for tripling frequency, the circuitcomprising: first and second input nodes configured to receive an inputvoltage having a sinusoidal shape and a base frequency; and first andsecond transistor pairs that are cross-coupled, the circuit having atrans-characteristics approximating a polynomial nominaltrans-characteristic defined by:${f\left( V_{in} \right)} = {\left( {{\frac{3}{A}V_{in}} - {\frac{4}{A^{3}}V_{in}^{3}}} \right)g_{m}}$wherein f(V_(in)) represents the trans-characteristic of the circuit,V_(in) represents the input voltage, A represents an amplitude of theinput voltage, and g_(m) is a transconductance of transistors of thefirst and second transistor pairs.
 2. The circuit of claim 1, furthercomprising third and fourth input nodes and first and second outputnodes, wherein: the first transistor pair comprises first and secondtransistors having respective first and second conduction terminals anda respective control terminal; the second transistor pair comprisesthird and fourth transistors having respective first and secondconduction terminals and a respective control terminal; the firstconduction terminals of the first, second, third and fourth transistorsare coupled together; the second conduction terminals of the first andfourth transistors are coupled together and to the first output node;the second conduction terminals of the second and third transistors arecoupled together and to the second output node; the control terminal ofthe first transistor is coupled to the first input node and isconfigured to receive a first fraction of the input voltage decreased byan offset voltage; the control terminal of the second transistor iscoupled to the second input node and configured to receive acounter-phase of the first fraction of the input voltage decreased bythe offset voltage; the control terminal of the third transistor iscoupled to the third input node and is configured to receive a secondfraction of the input voltage; and the control terminal of the fourthtransistor is coupled to the fourth input node and is configured toreceive a counter-phase of the second fraction of the input voltage. 3.The circuit of claim 2, wherein the first fraction is ½.
 4. The circuitof claim 2, wherein the second fraction is selected so that thetrans-characteristics of the circuit and the polynomial nominaltrans-characteristic have same slopes at zero crossings.
 5. The circuitof claim 2, wherein the second fraction is comprised between 0.1 and0.35.
 6. The circuit of claim 5, wherein the second fraction is 0.2. 7.The circuit of claim 2, wherein the offset voltage satisfies$\frac{2V_{os}}{\left( {1 - \alpha} \right)} = {\frac{\sqrt{3}}{2}A}$wherein V_(OS) represents the offset voltage, and α is an attenuationfactor based on the second fraction.
 8. The circuit of claim 2, furthercomprising an envelope detector coupled to the first and second inputnodes, the envelope detector configured to generate the offset voltage.9. The circuit of claim 8, further comprising: an input transformerhaving a primary winding coupled between first and second inputterminals and a secondary winding coupled between the first and secondinput nodes, the secondary winding having a central tap coupled to afirst output of the envelope detector; and a divider network coupledbetween the first and second input nodes and having a central tapcoupled to a second output of the envelope detector.
 10. The circuit ofclaim 9, wherein the divider network comprises a first branch comprisingfirst and second resistors and first and second capacitors.
 11. Thecircuit of claim 10, wherein: the divider network comprises a secondbranch comprising a third capacitor coupled between the third and fourthinput nodes; the first capacitor has a first terminal coupled to thefirst input node and a second terminal coupled to the first resistor andto the third input node; the second capacitor has a first terminalcoupled to the second input node (23) and a second terminal coupled tothe second resistor and to the fourth input node; and the first andsecond resistors are coupled together at the central tap of the dividernetwork.
 12. The circuit of claim 9, wherein the envelope detectorcomprises: an input differential pair coupled to the first and secondinput nodes; a first voltage generating network coupled between theinput differential pair and the first output of the envelope detector;and a second voltage generating network coupled to the second output ofthe envelope detector.
 13. The circuit of claim 2, wherein the first,second, third, and fourth transistors are bipolar transistors ormetal-oxide-semiconductor field-effect transistors.
 14. The circuit ofclaim 1, further comprising an LC network coupled to the first andsecond transistor pairs, the LC network comprising a tail inductor and ashunt capacitor coupled in series.
 15. A method for tripling frequency,the method comprising: receiving an input voltage having a sinusoidalshape and a base frequency; and applying the input voltage to a triplercircuit comprising first and second transistor pairs that arecross-coupled and having a trans-characteristics approximating apolynomial nominal trans-characteristic defined by:${f\left( V_{in} \right)} = {\left( {{\frac{3}{A}V_{in}} - {\frac{4}{A^{3}}V_{in}^{3}}} \right)g_{m}}$wherein f(V_(in)) represents the trans-characteristic of the circuit,V_(in) represents the input voltage, A represents an amplitude of theinput voltage, and g_(m) is a transconductance of transistors of thefirst and second transistor pairs; and generating an output currenthaving sinusoidal shape and an output frequency that is a triple of thebase frequency.
 16. The method of claim 15, wherein: the firsttransistor pair comprises a first and second transistors havingrespective first and second conduction terminals and a respectivecontrol terminal; the second transistor pair comprises third and fourthtransistors having respective first and second conduction terminals anda respective control terminal, the first conduction terminal of thefirst, second, third and fourth transistors are coupled to a commonnode; the second conduction terminal of the first and fourth transistorsare coupled to a first output node; the second conduction terminal ofthe second and third transistors are coupled to a second output node;applying the input voltage comprises: applying a first fraction of theinput voltage decreased by an offset voltage to the control terminal ofthe first transistor, applying a counter-phase of the first fraction ofthe input voltage decreased by the offset voltage to the controlterminal of the second transistor, applying a second fraction of theinput voltage to the control terminal of the third transistor, applyinga counter-phase of the second fraction of the input voltage to thecontrol terminal of the fourth transistor; and generating an outputcurrent comprises generating a first single-ended current at the firstoutput node and generating a second single-ended current at the secondoutput node.
 17. The method of claim 16, wherein the first fraction is ½and the second fraction is selected so that the trans-characteristics ofthe tripler circuit and the polynomial nominal trans-characteristic havesame slopes.
 18. The method of claim 16, wherein the second fraction iscomprised between 0.1 and 0.35.
 19. The method of claim 18, wherein thesecond fraction is 0.2.
 20. The method of claim 16, wherein the offsetvoltage satisfies condition:$\frac{2V_{os}}{\left( {1 - \alpha} \right)} = {\frac{\sqrt{3}}{2}A}$wherein V_(OS) represents the offset voltage, and a is an attenuationfactor based on the second fraction.
 21. The method of claim 16, furthercomprising detecting the amplitude of the input voltage and generatingthe offset voltage.
 22. A system comprising a frequency triplercomprising: first and second input nodes configured to receive an inputvoltage having a sinusoidal shape and a base frequency; third and fourthinput nodes; first and second output nodes; and first and second bipolartransistor pairs that are cross-coupled, the first bipolar transistorpair comprising first and second bipolar transistors having respectivefirst and second conduction terminals and a respective control terminal,the second bipolar transistor pair comprising third and fourth bipolartransistors having respective first and second conduction terminals anda respective control terminal, wherein the first conduction terminals ofthe first, second, third and fourth transistors are coupled together,the second conduction terminals of the first and fourth transistors arecoupled together and to the first output node, and the second conductionterminals of the second and third transistors are coupled together andto the second output node, wherein the frequency tripler has atrans-characteristics approximating a polynomial nominaltrans-characteristic defined by:${f\left( V_{in} \right)} = {\left( {{\frac{3}{A}V_{in}} - {\frac{4}{A^{3}}V_{in}^{3}}} \right)g_{m}}$wherein f(V_(in)) represents the trans-characteristic of the frequencytripler, V_(in) represents the input voltage, A represents an amplitudeof the input voltage, and g_(m) is a transconductance of bipolartransistors of the first and second transistor pairs.
 23. The system ofclaim 22, further comprising: a low-frequency voltage generatorconfigured to provide the input voltage to the frequency tripler; and afrequency doubler configured to generate an output voltage based on anoutput of the frequency tripler.